Enhanced DC offset mitigation

ABSTRACT

A technique is disclosed to estimate the frequency offset in received direct downconverted predetermined signal despite the presence of a DC offset component. After compensation of the received predetermined signal according to the frequency offset estimation, the DC offset component is estimated. DC offset in data signals subsequent to the predetermined signal may then be mitigated based upon the estimated DC offset component.

FIELD OF INVENTION

[0001] This invention relates to direct down conversion and morespecifically to mitigating the DC offset component in adirect-down-converted signal.

BACKGROUND

[0002] Radio receivers typically employ a superheterodyne architecture.In this architecture, the receiver uses two stages to translate an RFsignal from a carrier frequency to an intermediate frequency (IF) andthen to baseband. In contrast, a direct downconversion receiver usesjust one stage to translate an RF signal directly from the carrierfrequency to baseband. Because the extra stage in a superheterodynereceiver inevitably introduces additional noise (thereby degrading thesignal-to-noise ratio) and requires more components (making productionmore costly), direct downconversion receivers pose an attractivealternative to the prevalent superheterodyne architecture.

[0003] In a direct downconversion receiver, the RF signal is mixed witha local oscillator (LO) signal. Inevitably, this process produces anundesired self mixing of the LO signal, thereby producing both a DCoffset component and a high frequency component. For example, if the LOsignal is represented by the sinusoid cos(ωt), the self mixing producesthe product:

cos(ωt)*cos(ωt)=½(the DC offset component)+cos(2ωt)/2 (the highfrequency component).

[0004] The high frequency component may be filtered off. However, the DCoffset component can wreak havoc in subsequent baseband processing,particularly for higher throughput modulations. For example, FIG. 1shows the potential locations 10 in signal space for a QPSK-modulatedsignal. Signals 12 show the subsequent location in signal space shouldan arbitrary DC offset component be present. Although distorted, signals12 may still be correctly demodulated. But should the same DC offsetcomponent be present for signals 14 in a 16-QAM modulation scheme asshown by resulting signals 16 in FIG. 2, errors may result as indicatedby locations 16 a. Accordingly, the presence of a DC offset poses aserious problem for any direct-downconverted receiver architecture. Asuperheterodyne receiver, however, may filter off the DC offsetcomponents because of the extra stage of processing. Thus, despite theirinferior noise properties and higher manufacturing costs,superheterodyne receivers are more popular than direct downconversionreceivers.

[0005] To realize the benefits of a direct downconversion receiver,something must be done to mitigate the DC offset component. For example,the DC offset component may be estimated so that it may be subsequentlycompensated for at baseband. But the measurement of the DC offsetcomponent becomes problematic should a frequency offset be presentbetween the receiver and the transmitter. Such a frequency offset isinherent in any communication system, particularly in mobileapplications subject to Doppler effects.

[0006] Accordingly, there is a need in the art for improved techniquesto estimate the DC offset component despite the presence of a frequencyoffset between the receiver and transmitter.

SUMMARY

[0007] In accordance with one aspect of the invention, a method ofmitigating a DC offset component in a received data sample subject to afrequency offset includes an act of receiving a sequence of samples of apredetermined signal, wherein the predetermined signal is periodic overn samples. Because the predetermined signal is periodic with respect ton samples, the frequency-offset-induced phase shift between successivereceived samples of the predetermined signal may be estimated bycomparing received samples in a first period of the predetermined signalto the corresponding received samples in a second period of thepredetermined signal. The DC offset component in the received samples ofthe predetermined signal may then be estimated after compensating thereceived samples based upon the estimated frequency-offset-induced phaseshift. This estimated DC offset component may then be cancelled for inthe received data sample.

[0008] In accordance with another aspect of the invention, a basebandprocessor includes a state machine for estimating afrequency-offset-induced phase shift between successive received samplesof a predetermined signal, wherein the transmitted samples of thepredetermined signal are periodic. The state machine estimates thisphase shift by comparing received samples in a first period of thepredetermined signal to the corresponding received samples in a secondperiod of the predetermined signal. The state machine is furtherconfigured to estimate a DC offset component in the received samples ofthe predetermined signal after compensating the received samplesaccording to the estimated phase shift.

[0009] The invention will be more fully understood upon consideration ofthe following detailed description, taken together with the accompanyingdrawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0010]FIG. 1 illustrates the effect of a DC offset component on thepotential locations in signal space for a QPSK-modulated signal.

[0011]FIG. 2 illustrates the effect of a DC offset component on thepotential locations in signal space for a 16-QAM-modulated signal.

[0012]FIG. 3 is a z-transform representation of a prior art averagingtechnique that estimates the DC component without accounting forfrequency offsets.

[0013]FIG. 4 is a shift register implementation for the techniqueillustrated in FIG. 3.

[0014]FIG. 5 is a z-transform representation of a circuit to estimatethe DC offset component in the presence of a frequency offset accordingto one embodiment of the invention.

[0015]FIG. 6 is a partial block diagram for a DC offset cancellationprocessor according to one embodiment of the invention.

[0016]FIG. 7 is a z-transform implementation of a frequency offsetestimation technique that does not compensate for the presence of a DCoffset component according to one embodiment of the invention.

[0017]FIG. 8 is a z-transform implementation of a frequency offsetestimation technique independent of the presence of a DC offsetcomponent according to one embodiment of the invention.

[0018]FIG. 9 is a block diagram for a DC offset cancellation processoraccording to one embodiment of the invention.

DETAILED DESCRIPTION

[0019] In networks such as a wireless LAN, data transmission occurs inbursts or packets of data. Each burst of data may be pre-pended with apreamble that is known to the receiver. Because the characteristics ofthis preamble are known apriori to the receiver, it may be used forsignal up detection, automatic gain control (AGC), antenna diversityimplementation, and receiver synchronization. As will be explainedfurther herein, the known characteristics of the preamble may also beexploited by the receiver to aid in DC offset component estimation.

[0020] For example, in an IEEE 802.11a waveform, the preamble consistsof ten identical sequences each of length sixteen. Each sequence has anaverage value of zero. Thus, by summing over all sixteen samples in eachsequence, a DC offset component may be estimated. Mathematically, thekth received sample {tilde over (r)}_(k) may be given by:

{tilde over (r)} _(k) ={tilde over (p)} _(k) +{tilde over (V)} _(DC)

[0021] where {tilde over (V)}_(DC) is the DC offset component for eachreceived sample and {tilde over (p)}_(k) is the transmitted kth sampleof the preamble. Those of ordinary skill will appreciate that thesesignals may be complex or real depending upon the signal representation.It follows that:${\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{r}}_{k}} = {{\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{p}}_{k}} + {n{\overset{\sim}{V}}_{D\quad C}}}$

[0022] where n is the number of samples for summation. A typical valueof n for an IEEE 802.11a application would be 16 or 32. Afterrearrangement, the previous equation becomes:${\overset{\sim}{V}}_{D\quad C} = \frac{{\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{r}}_{k}} - {\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{p}}_{k}}}{n}$

[0023] The known periodic characteristics of the preamble may be used tosolve for the DC component. Assuming an IEEE 802.11a application, theaverage value of the n samples in a complete sequence from a preamble iszero such that:${\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{p}}_{k}} = 0$

[0024] resulting in the following expression for {tilde over (V)}_(DC):${\overset{\sim}{V}}_{D\quad C} = {\frac{1}{n}{\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{r}}_{k}}}$

[0025]FIG. 3 illustrates a z-transform implementation of the previousderivation for the DC offset component. It will be appreciated that avariety of circuits may be used to implement the z-transformrepresentation of FIG. 3. For example, FIG. 4 shows an averaging circuit14 that estimates {tilde over (V)}_(DC) using a 16-bit shift register 20and a D-type flip-flop 22. Averaging circuit 14 may be implemented forboth the in-phase (I) and the quadrature-phase (Q) signal components.

[0026] The averaging approach discussed with respect to FIGS. 3 and 4becomes problematic, however, should a frequency offset be present inthe received signal from a Doppler shift or other effects. Because thefrequency offset is not known, the characteristics of the receivedsignal cannot be predicted apriori despite the known characteristics ofthe preamble. For example, let ω_(o) be the frequency offset and ω_(s)be the sampling frequency. The received signal then becomes:

{tilde over (r)} _(k) ={tilde over (p)} _(k) e ^(jφ) ^(_(k)) +{tildeover (V)} _(DC)  Equation (1)

[0027] where φ_(k)=k−φ_(o) and φ_(o)=. Summing all received signalsamples over n gives:${\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{r}}_{k}} = {{\sum\limits_{k = 0}^{n - 1}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}} + {n\quad {\overset{\sim}{V}}_{D\quad C}}}$

[0028] Solving for {tilde over (V)}_(DC) in the previous equation gives:${\overset{\sim}{V}}_{D\quad C} = {\frac{1}{n}\left( {{\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{r}}_{k}} - {\sum\limits_{k = 0}^{n - 1}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}}} \right)}$

[0029] where the second term in the parenthesis is not zero if afrequency offset exists.

[0030] Accordingly, the performance of communication systems thatestimate the DC offset component using the averaging approach of FIG. 3is not very encouraging should even a small frequency offset be present.The frequency offset should be removed before a reliable DC offsetestimation can be performed. The present invention provides twotechniques to estimate and remove the frequency offset. In a firsttechnique, the frequency offset is estimated in the presence of the DCoffset component. In a second technique, the estimate from the firsttechnique is refined by removing the influence of the DC offsetcomponent. Regardless of the estimation technique implemented, thefollowing approach may be used to mitigate the DC offset component inthe presence of a frequency offset in a direct-down converted signalsuch as that produced by the receiver architecture disclosed in theco-pending U.S. pat. application Ser. No. ______, entitled “ZeroIntermediate Frequency to Low Intermediate Frequency ReceiverArchitecture,” Attorney docket no. M-15027, filed ______, 2002, thecontents of which are hereby incorporated by reference.

[0031] If the estimated frequency offset is represented by {circumflexover (ω)}_(o), the estimated amount of frequency-offset-induced phaseshift between each received sample of the preamble would be${\hat{\varphi}}_{o} = {2\quad \pi \quad {\frac{{\hat{\omega}}_{o}}{\omega_{s}}.}}$

[0032] From equation (1) and the estimated amount of phase shift{circumflex over (φ)}_(o) in each received sample, the followingestimate for {tilde over (V)}_(DC) may be derived:

{tilde over (r)} _(k) e ^(−jk{circumflex over (φ)}) ^(_(o)) ={tilde over(p)} _(k) e ^(jφ) ^(_(k)) e ^(−jk{circumflex over (φ)}) ^(_(o)) +{tildeover (V)} _(DC) e ^(−jk{circumflex over (φ)}) ^(_(o))

{tilde over (r)} _(k) e ^(−jk{circumflex over (φ)}) ^(_(o)) ={tilde over(p)} _(k) e ^(j(φ) ^(_(k)) ^(−k{circumflex over (φ)}) ^(_(o)) ⁾ +{tildeover (V)} _(DC) e ^(−jk{circumflex over (φ)}) ^(_(o)) $\begin{matrix}{{\sum\limits_{k}{{\overset{\sim}{r}}_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}} = {{\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j{({\varphi_{k} - {k{\hat{\varphi}}_{o}}})}}}} + {{\overset{\sim}{V}}_{D\quad C}{\sum\limits_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}}}} \\{{\overset{\sim}{V}}_{D\quad C} = \frac{{\sum\limits_{k}{{\overset{\sim}{r}}_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}} - {\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j{({\varphi_{k} - {k\quad {\hat{\varphi}}_{o}}})}}}}}{\sum\limits_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}}\end{matrix}$

[0033] Assuming that the frequency estimate is accurate, the quantity(φ_(k)−k{circumflex over (φ)}_(o)) would be very small, such that the DCoffset is given by: $\begin{matrix}{{\overset{\sim}{V}}_{D\quad C} = \frac{{\sum\limits_{k}{{\overset{\sim}{r}}_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}} - {\sum\limits_{k}{\overset{\sim}{p}}_{k}}}{\sum\limits_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}} & {{Equation}\quad (2)}\end{matrix}$

[0034] For an IEEE 802.11a waveform, the second term in the numerator ofEquation (2) is very small or zero and can be neglected. FIG. 5illustrates a z-transform implementation of Equation (2). In frequencyestimation block 40, {circumflex over (φ)}_(o) is derived according tothe present invention as discussed herein. Although the approach of FIG.5 is feasible, the required division would entail considerablecomplexity when implemented in hardware. It will be appreciated that theDC offset estimation techniques disclosed herein may be implemented ineither hardware or software. A software approach would avoid thecomplexities of the hardware division required to implement the DCoffset estimation discussed with respect to FIG. 5. However, dedicatedstate machines implemented, e.g., in an ASIC, will typically providegreater processing speed.

[0035] Because a dedicated hardware approach is desirable but, ifimplemented based upon the z-transform representation of FIG. 5, wouldinvolve considerable complexity, an alternate approach is as follows. ADC offset mitigation using the estimate provided by the Equation (2)would be performed on samples {tilde over (r)}_(t) of data transmittedafter the known samples of the preamble such that a corrected datasample Out_(a) is given by: $\begin{matrix}{{Out}_{a} = {{{\overset{\sim}{r}}_{t} - {\overset{\sim}{V}}_{D\quad C}} = {{\overset{\sim}{r}}_{t} - \frac{{\sum\limits_{k}{{\overset{\sim}{r}}_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}} - {\sum\limits_{k}{\overset{\sim}{p}}_{k}}}{\sum\limits_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}}}} & {{Equation}\quad (3)}\end{matrix}$

[0036] Multiplication of Equation (3) with the numerator term$\left( {\sum\limits_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}} \right)$

[0037] gives the following corrected data sample Out_(b):$\begin{matrix}\begin{matrix}{{Out}_{b} = {{{\overset{\sim}{r}}_{t}{\sum\limits_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}} - \left( {{\sum\limits_{k}{{\overset{\sim}{r}}_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}}} - {\sum\limits_{k}{\overset{\sim}{p}}_{k}}} \right)}} \\{= {{Out}_{a}\left( {\sum\limits_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}} \right)}}\end{matrix} & {{Equation}\quad (4)}\end{matrix}$

[0038] This factor$\left( {\sum\limits_{k}^{{- j}\quad k\quad {\hat{\varphi}}_{o}}} \right)$

[0039] generates a known amplitude and phase change, which may becompensated for in subsequent demodulation to give the desired correcteddata sample value Out_(a) described previously. FIG. 6 illustrates a DCoffset mitigation processor 60 performing the DC offset cancellation ofEquation (4). Advantageously, no complex hardware is required—those ofordinary skill in the art will appreciate the numerous ways such a DCoffset mitigation processor may be implemented using simple hardwaresuch as shift registers, multipliers, and adders. Alternatively, such aDC offset mitigation processor may be performed using a microprocessorin a software-based approach. The techniques of the present invention toprovide the frequency offset estimation factor {circumflex over (φ)}_(o)will now be discussed.

[0040] Frequency Offset Estimation Techniques

[0041] 1. Estimating Frequency Offset with the DC Offset Component.

[0042] As can be seen from Equation (1), the kth received preamblesample {tilde over (r)}_(k) will be shifted in phase by an amount(k*{circumflex over (φ)}_(o)) with respect to the kth transmitted sample{tilde over (p)}_(k) of the preamble (ignoring, for the moment, thecontribution from {tilde over (V)}_(DC)). Assuming the preamblecomprises at least two identical sequences each having n samples, thefollowing technique may be used. At the (k+n)th received sample of thepreamble, the amount of phase shift will be (n+k)*{circumflex over(φ)}_(o) with respect to the (k+n)th transmitted sample in the preamble.But given the periodicity over n of the preamble, the (k+n)thtransmitted sample is the same as the kth transmitted sample.Accordingly, if the (k+n)th received sample is multiplied by the complexconjugate of the kth received sample, the product is the phasorexp(j{circumflex over (φ)}_(o)*n) (ignoring the amplitude component,which has no effect on the phase). The frequency offset per sample{circumflex over (φ)}_(o) may thus be derived by taking the arc tangentof this phasor and dividing the result by n. However, this discussionignores the contribution from the DC offset component. But for a smallDC offset component, the estimation is quite good.

[0043] A z-transform implementation of the preceding algorithm is shownin FIG. 7, wherein the periodicity of the transmitted samples isrepresented by the variable “m.” To decrease noise, the kth receivedsample is multiplied by the complex conjugate of the (k−m)th receivedsample for all samples in the sequence (from k=0 to k=(m−1)) and thensummed. Alternatively, greater or fewer samples could be summeddepending upon the desired signal-to-noise ratio and latencyrequirements (where decreasing the number of samples would decreaselatency but also decrease the signal-to-noise ratio). Although the DCoffset component is estimated with this baseline frequency offset, thecomputation is considerably better than the simple averaging approachdiscussed previously—there are no error floors and the performance isonly slightly degraded with the expected DC offset.

[0044] 2. Estimating Frequency Offset Independently of the DC OffsetComponent.

[0045] In a second approach, an enhanced frequency offset is estimatedindependently from any effects of a DC offset component present in thesystem. The following is a derivation of this approach. As discussedpreviously, the kth received sample {tilde over (r)}_(k) of the preamblemay be represented by:

{tilde over (r)} _(k) ={tilde over (p)} _(k) e ^(jφ) ^(_(k)) +{tildeover (V)} _(DC)  Equation (4)

[0046] Similarly, the (k−m)th sample of the preamble, where m is thenumber of samples in a sequence in the preamble is given by:

{tilde over (r)} _(k−m) ={tilde over (p)} _(k−m) e ^(jφ) ^(_(k−m))+{tilde over (V)} _(DC) ={tilde over (p)} _(k) e ^(jφ) ^(_(k)) e ^(−jmφ)^(_(o)) +{tilde over (V)} _(DC)

[0047] it follows that:

{tilde over (r)} _(k−m) e ^(jmφ) ^(_(o)) ={tilde over (p)} _(k) e ^(jφ)^(_(k)) +{tilde over (V)} _(DC) e ^(jmφ) ^(_(o))   Equation (5)

[0048] From equations (4) and (5) it can be shown that: $\begin{matrix}\begin{matrix}{{\sum\limits_{k}{{\overset{\sim}{r}}_{k}{\overset{\sim}{r}}_{k - m}^{*}}} = {{^{j\quad m\quad \varphi_{o}}{\sum\limits_{k}{{\overset{\sim}{p}}_{k}}^{2}}} + {m{{\overset{\sim}{V}}_{D\quad C}}^{2}} +}} \\{{{^{j\quad m\quad \varphi_{o}}{{\overset{\sim}{V}}_{D\quad C}\left( {\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}} \right)}^{*}} + {{\overset{\sim}{V}}_{D\quad C}^{*}\left( {\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}} \right)}}}\end{matrix} \\{and} \\\begin{matrix}{{\left( {\sum\limits_{k}{\overset{\sim}{r}}_{k}} \right)\left( {\sum\limits_{k}{\overset{\sim}{r}}_{k - m}} \right)^{*}} = {{^{j\quad m\quad \varphi_{o}}{{\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}}}^{2}} + {{m\quad {\overset{\sim}{V}}_{D\quad C}}}^{2} +}} \\{{{m\quad ^{j\quad m\quad \varphi_{o}}{{\overset{\sim}{V}}_{D\quad C}\left( {\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}} \right)}^{*}} + {m{{\overset{\sim}{V}}_{D\quad C}^{*}\left( {\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}} \right)}}}}\end{matrix} \\{{such}\quad {{that}:}} \\{{{\sum\limits_{k}{{\overset{\sim}{r}}_{k}{\overset{\sim}{r}}_{k - m}^{*}}} - {\frac{1}{m}\left( {\sum\limits_{k}{\overset{\sim}{r}}_{k}} \right)\left( {\sum\limits_{k}{\overset{\sim}{r}}_{k - m}} \right)^{*}}} = {{^{j\quad m\quad \varphi_{o}}{\sum\limits_{k}{{\overset{\sim}{p}}_{k}}^{2}}} - {\frac{1}{m}^{j\quad m\quad \varphi_{o}}{{\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}}}^{2}}}}\end{matrix}$

[0049] The preceding equation may be factored as follows:${{\sum\limits_{k}{{\overset{\sim}{r}}_{k}{\overset{\sim}{r}}_{k - m}^{*}}} - {\frac{1}{m}\left( {\sum\limits_{k}{\overset{\sim}{r}}_{k}} \right)\left( {\sum\limits_{k}{\overset{\sim}{r}}_{k - m}} \right)^{*}}} = {^{j\quad m\quad \varphi_{o}}\left( {{\sum\limits_{k}{{\overset{\sim}{p}}_{k}}^{2}} - {\frac{1}{m}{{\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}}}^{2}}} \right)}$

[0050] Because the factor$\left( {{\sum\limits_{k}{{\overset{\sim}{p}}_{k}}^{2}} - {\frac{1}{m}{{\sum\limits_{k}{{\overset{\sim}{p}}_{k}^{j\quad \varphi_{k}}}}}^{2}}} \right)$

[0051] is a real number, the estimated phase shift {circumflex over(φ)}_(o) per received preamble sample from this enhanced technique isthus given by:$\varphi_{o} = {\frac{1}{m}{phase}\left\{ {{\sum\limits_{k}{{\overset{\sim}{r}}_{k}{\overset{\sim}{r}}_{k - m}^{*}}} - {\frac{1}{m}\left( {\sum\limits_{k}{\overset{\sim}{r}}_{k}} \right)\left( {\sum\limits_{k}{\overset{\sim}{r}}_{k - m}} \right)^{*}}} \right\}}$

[0052]FIG. 8 is the z-transform implementation of this enhancedfrequency offset estimate that is independent of effects from the DCoffset component. Those of ordinary skill in the art will appreciatethat a number of different state machines may be designed to implementthis enhanced frequency offset estimation technique. For example, delayblock 70, for both the I and Q components, may be implemented usingshift register 20 and flip-flop 22 discussed with respect to FIG. 4.Similar delay blocks also required for this technique may be implementedin that fashion as well.

[0053] Regardless of how the frequency offset is estimated, the presentinvention provides an enhanced DC offset mitigation technique that isrobust with respect to frequency offset effects. FIG. 9 illustrates areceiver 100 configured to perform this DC offset mitigation technique.A direct downconversion receiver 110 receives an RF signal and providesan analog baseband signal to an analog-to-digital converter 120.Analog-to-digital converter 120 digitizes the analog baseband signal andprovides the resulting received signal samples to a frequency offsetestimation state machine 130. State machine 130 may perform eitherfrequency offset technique discussed herein to provide the estimatedfrequency-offset-induced phase shift {circumflex over (φ)}_(o) betweeneach received sample of the preamble. DC cancellation state machine 140receives the digitized baseband samples and the phase shift estimationfactor {circumflex over (φ)}_(o) to cancel the DC offset componentaccording to the techniques disclosed herein. This cancellation mayinvolve the complex division discussed with respect to FIG. 5 or beperformed as discussed with respect to FIG. 6. In an alternateembodiment, state machines 130 and 140 may be combined.

[0054] Although the present invention has been described with respect toan IEEE 802.11a based communication system, it may be applied to anycommunication scheme that transmits data that is prepended with apredetermined periodic signal. Because the properties of thispredetermined periodic signal are known apriori by the receiver, thereceived samples from this predetermined packet may be exploitedaccording to the present invention to mitigate the DC offset componentdespite the presence of a frequency offset between the transmitter andthe receiver. For example, if the predetermined signal is periodic overn samples, a given sample of the predetermined signal should have thesame phase as a sample delayed with respect to the given sample by nsamples. By examining the phase shift between these two samples, thefrequency offset estimation may be performed according to the techniquesdisclosed herein. Using this frequency offset estimate and the knownvalues of the transmitted predetermined signal samples, the DC offsetcomponent for the received predetermined signal samples may also bederived according to the techniques disclosed herein. This DC offsetcomponent may then be cancelled in any data samples received subsequentto the predetermined signal. Accordingly, although the invention hasbeen described with respect to particular embodiments, this descriptionis only an example of the invention's application and should not betaken as a limitation. Consequently, the scope of the invention is setforth in the following claims.

What is claimed is:
 1. A method of mitigating a DC offset component,comprising: (a) estimating a frequency-offset-induced phase shift{circumflex over (φ)}_(o) between successive received samples of apredetermined signal by comparing at least one sample in a first periodof the predetermined signal with the corresponding at least one receivedsample in a second period of the predetermined signal; (b) estimating aDC offset component in the received samples of the predetermined signalafter compensating the received samples according to the estimated phaseshift {circumflex over (φ)}_(o); and (c) mitigating a DC offsetcomponent in a received data sample according to the estimated DCoffset.
 2. The method of claim 1, wherein the first and second period ofthe predetermined signal each comprises n samples, and wherein act (c)comprises: (e) forming the sum of the n successive received samples ofthe a period of the predetermined signal, each received sample in thesum being rotated in phase according to the estimated phase shift{circumflex over (φ)}_(o) such that a jth received sample is rotated inphase by an amount −j times the phase shift {circumflex over (φ)}_(o);(f) subtracting the sum of n transmitted samples for a period of thepredetermined signals from the result from act (e); and (g) adjustingthe received data sample using the result from act (f).
 3. The method ofclaim 2, wherein act (g) comprises: (i) forming the sum:$\sum\limits_{k = 0}^{n - 1}^{- {j{({k\quad {\hat{\varphi}}_{o}})}}}$

(j) dividing the result from act (f) with the result from act (i); and(k) subtracting the result from act (j) from the received data sample.4. The method of claim 2, wherein act (g) comprises: (i) multiplying thereceived data sample by the sum:$\sum\limits_{k = 0}^{n - 1}^{- {j{({k\quad {\hat{\varphi}}_{o}})}}}$

(j) subtracting the result from act (f) from the result from act (i). 5.The method of claim 1, wherein the estimation in act (a) is made withoutcompensating for a DC offset component in the received samples of thepredetermined signal.
 6. The method of claim 5, wherein act (a)comprises extracting the phase difference between the at least onereceived sample in the first period and the corresponding at least onereceived sample in the second period.
 7. The method of claim 5, whereinact (a) comprises averaging the phase difference between each receivedsample in the first period and the corresponding received sample in thesecond period.
 8. The method of claim 7, wherein the predeterminedsignal is a preamble of an IEEE 802.11a modulated data signal.
 9. Themethod of claim 5, wherein the estimation in act (a) is madeindependently from a DC offset component in the received samples of thepredetermined signal.
 10. The method of claim 9, wherein act (a)comprises: for each given sample {tilde over (r)}_(k) in the firstperiod and the corresponding sample {tilde over (r)}*_(k−n) in thesecond period, calculating {tilde over (φ)}_(o) such that:${\hat{\varphi}}_{o} = {{{phase}\left( {{\sum\limits_{k = 0}^{n - 1}\left\{ {{\overset{\sim}{r}}_{k}{\overset{\sim}{r}}_{k - n}^{*}} \right\}} - {\frac{1}{n}\left( {\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{r}}_{k}} \right)\left( {\sum\limits_{k = 0}^{n - 1}{\overset{\sim}{r}}_{k - n}} \right)^{*}}} \right)}.}$


11. The method of claim 10, wherein the predetermined signal is apreamble of an IEEE 802.11a modulated data signal.
 12. A basebandprocessor for a received direct downconverted predetermined signal,wherein the received predetermined signal is subject to a frequencyoffset, comprising: an analog-to-digital converter for digitizingsamples of the received predetermined signal; and a state machineconfigured to estimate a frequency-offset-induced phase shift{circumflex over (φ)}_(o) between successive received samples of thepredetermined signal by comparing at least one sample in a first periodof the predetermined signal with the corresponding at least one receivedsample in a second period of the predetermined signal, wherein the statemachine is further configured to estimate a DC offset component in thereceived samples of the predetermined signal after compensating thereceived samples according to the estimated phase shift {circumflex over(φ)}_(o).
 13. The baseband processor of claim 12, wherein the statemachine comprises an application-specific-integrated circuit (ASIC). 14.The baseband processor of claim 12, wherein the state machine isconfigured to estimate {circumflex over (φ)}_(o) without compensatingfor a DC offset component in the received samples of the predeterminedsignal.
 15. The baseband processor of claim 12, wherein the statemachine is configured to estimate {circumflex over (φ)}_(o)independently from a DC offset component in the received samples of thepredetermined signal.
 16. The baseband processor of claim 14, whereinthe predetermined signal is a preamble of an IEEE 802.11a modulated datasignal.
 17. The baseband processor of claim 15, wherein thepredetermined signal is a preamble of an IEEE 802.11a modulated datasignal.